Method and apparatus for transmission line equalization

ABSTRACT

A method and apparatus for equalizing the frequency response of a transmission line is provided. The method includes the steps of modelling the frequency response of the transmission media for a predetermined frequency range to a predetermined accuracy; determining a desired equalizer response by taking an inverse of the modelled frequency response of the first step; implementing an equalizer that exhibits the desired response; and utilizing the equalizer to equalize the frequency response of the transmission line. The apparatus includes an adaptive equalizer circuit which includes a plurality of signal processor circuits which each take an input signal from the transmission line and process it to mimic a term in a transfer function which represents an inverse of the transfer function of the transmission line. The signals from these processors are then summed and multiplied by a programmable gain term. Then the input is added to the output of the multiplier to form an output equalizer signal. The programmable gain term is adapted to cause the signal at the output to closely approximate a predetermined high frequency signal peak value through a send loop mechanism.

CROSS REFERENCE TO A RELATED APPLICATION

This application is a divisional of application Ser. No. 08/699,031,filed Aug. 15, 1996, still pending.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to the field of data communications systems. Morespecifically, it relates to the field of wired communications systems.Even more specifically, it relates to a method and apparatus forproviding compensation or equalization for the frequency response of atransmission line.

2. The Prior Art

At some point in any data communications system, the signal to becommunicated passes through an electrical conductor. This electricalconductor can take many forms such as, for example, a coaxial cable("coax"), an unshielded twisted pair cable ("UTP"), or a shieldedtwisted pair cable ("STP"). Such conductors are known generally astransmission media or transmission lines. Most transmission lines suchas these exhibit a low pass characteristic. That is, they transmit lowfrequency components of the signal more readily than high frequencycomponents of the signal, i.e., their frequency response is not "flat".

Most communications signals consist of symbols that represent theinformation to be communicated. These symbols are usually packed closeto one another in the time domain in order to achieve the highesttransmission speeds. However, when a signal such as this is passedthrough a transmission line having other than a flat frequency response,the low pass characteristic of the transmission line has the effect ofwidening each symbol in time. This widening can result in spill over orinter-symbol interference among symbols of the signal. This in turn canresult in the loss of, or incorrect communication of information.

To compensate for the low pass characteristic of a transmission line,the signal is typically passed through a transmission line equalizer atthe receiver end of the transmission line. The transmission lineequalizer exhibits a high pass characteristic. That is, it transmitshigh frequency components of the signal more readily than the lowfrequency components of the signal and therefore exhibits an inversefrequency response to that of the transmission line. When put in serieswith the low pass frequency response of the transmission line, the highpass frequency response of the transmission line equalizer has theeffect of returning each symbol to its original form. The information tobe communicated is thereby preserved.

The exact low pass characteristic of the transmission line depends inpart on the specific media used and in part on the length of thetransmission line. So, if these factors vary within a particularapplication of a communications system, the transmission line equalizerwill have to be able to adapt to these differences in order to correctlycompensate for the effect of the transmission line on the signal.

It is important that the transmission line equalizer provides just theright amount of compensation to avoid under-compensation (the failure toremove some residual inter-symbol interference that continues to causethe loss of or incorrect communication of information) orover-compensation (noise enhancement and distortion of the symbols thatcauses the loss of or incorrect communication of information).

As those of ordinary skill in the art will recognize, the frequencyresponse of any system can be expressed by a polynomial equationreferred to as the transfer function of the system. This equation oftentakes the shape of a fraction with the numerator having one polynomialexpression and the denominator having another. The roots of thenumerator are referred to as zeros and the roots of the denominator arereferred to as poles. These roots correspond to the corner frequenciesof the system and can be manipulated by a designer to create a transferfunction having a desired frequency response.

In the case of a communications system, the transmission line exhibitsone transfer function and the transmission line equalizer attempts toexhibit the exact inverse of that transfer function in order tocompensate for the effect of the transmission line on the communicatedsignal.

In order to achieve successful compensation, two general steps must beaccomplished. First, the frequency response of the transmission linemust be either determined empirically (i.e., by measurement) orapproximated theoretically. Since empirical determination is often costprohibitive, theoretical approximation is generally the method used. Theaccuracy of such an approximation will depend in part on the specificmedia used and the length of the transmission line. Second, the inverseof the frequency response of the transmission line must be exhibited bythe transmission line equalizer as configured by the designer. This isaccomplished by manipulating the number and location of the roots, polesand zeros of the transfer function of the transmission line equalizeruntil the desired frequency response is achieved. As noted above, theaccuracy of the approximation of the frequency response of thetransmission line depends on the specific media used and the length ofthe transmission line. Hence, the transmission line equalizer should bedesigned to adapt to changes in either of these factors to assuresuccessful compensation for the signal.

The frequency response of a UTP or STP transmission line due to ordominated by skin effect is usually approximated as having a roll off of10 dB per decade (10 db/dec).

As discussed before, using this approximation a transmission lineequalizer would have to exhibit the inverse frequency response of thetransmission line to compensate correctly for the presumed frequencyresponse of the transmission line. Unfortunately, each zero in atransfer function provides a gain of 20 dB/dec above the cornerfrequency of that zero. This is obviously too much by itself. What isneeded is a transfer function with about "half" of the gain of one zero,i.e. 10 dB instead of 20 dB. The prior art has adopted two approaches toreduce this response to more closely approximate the inverse frequencyresponse of the typical transmission line.

The first prior art approach is to design the transmission lineequalizer with a pole/zero pair and place the pole at a higher frequencythan the zero. The addition of the pole has the effect of partiallycanceling the gain of the zero. The intended result is for the modifiedtransfer function to have a slope of 10 dB/dec in the frequency range ofinterest (i.e., the frequency range of the signals to be transmittedover the transmission line).

For an adaptive design, the frequency separation between the pole andzero can be varied to adjust the frequency response of the transmissionline equalizer to better match that of the transmission line whateverthe specific media used or the length of the transmission line.

FIG. 1A is a representation of a transfer function H(s) where srepresents the complex frequency, z is the zero frequency (rad/sec) ofthe transfer function, and p is the pole frequency (rad/sec) of thetransfer function where z is less than p.

FIG. 1B is a plot showing the frequency response of the transferfunction H(s) of FIG. 1A. The vertical axis is gain in db (20loglH(jω)l), the horizontal axis is angular frequency, ω, in radians persecond. The dashed curve is an asymptote of the transfer function H(s).The solid curve is a plot of the transfer function H(s).

The second prior art approach is to add an "all pass" function to a highpass function. An "all pass" function transmits all frequency componentsof the signal. The addition of the all pass function to the high passfunction has the effect of producing a weighted average of the twofunctions with less high frequency gain than that of the high passfunction alone.

The high pass function can have any number of zeros. Recall that onezero produces a gain of 20 dB/dec. This is true of each zero in thefunction. So at frequencies above the highest corner frequency, a highpass function that has n zeros would exhibit a gain of n x 20 dB/dec.

For an adaptive design, the mixture in the addition function can bevaried to adjust the frequency response of the transmission lineequalizer to better match that of the transmission media whatever thespecific media used or the length of that media. One technique used tovary the mixture is to multiply the output of the high pass function bya selectable constant.

FIG. 2A is a representation of a transfer function E(s) where srepresents the complex frequency. H₁ (s) is a high pass transferfunction (shown as a second order function specifically here), ω_(n) isthe natural frequency of H₁ (s), ξ is the damping factor of H₁ (s), k isthe variable gain (between 0 and 1) fitted to the transmission line typeand length.

FIG. 2B is a plot of gain versus frequency for four curves A, B, C andD. Curve A is a constant unity gain function. Curve B is the frequencyresponse of H₁ (s). Curves C and D are two different equalizer responsecurves corresponding to gain value k₁ and k₂, respectively, where k₁>k₂.

As can be seen, adjusting k₁ in transfer function E(s) between 0 and 1allows the slopes of curves C and D to be adjusted as desired.

The transmission line equalizers that result from either of the twoprior art approaches above can only provide a moderate approximation ofthe transfer function needed to compensate for the frequency response ofthe transmission lines in certain specific applications. This is due toinaccuracies in the model used to approximate the frequency response ofthe transmission media.

One specific application where the prior art transmission lineequalizers provide only a rough approximation of the transfer functionneeded to compensate for the frequency response of the transmissionmedia is for Fast Ethernet Local Area Network (LAN) applicationsreferred to as 100BaseT4 designed to IEEE Standard 802.3u-1995 Clause23. This is due in part to the wide frequency spread and in part to theadvanced coding scheme of the 100BaseT4 standard. The resulting mismatchbetween the transmission line and the transmission line equalizerdegrades the receiver performance to such an extent that it limits theachievable transmission speed and distance.

OBJECTS AND ADVANTAGES OF THE INVENTION

Accordingly, it is an object and advantage of the present invention toprovide a transmission line model that more closely approximates thefrequency response of actual transmission lines.

It is a further object and advantage of the present invention to providea transmission line equalizer that more closely exhibits a transferfunction that compensates for the frequency response of a transmissionline.

It is a further object and advantage of the present invention to providea transmission line equalization circuit which adaptively compensatesfor the frequency response of a transmission line.

Yet a further object and advantage of the present invention is toprovide an improved method of compensating for the frequency response ofa transmission line.

These and many other objects and advantages of the present inventionwill become apparent to those of ordinary skill in the art from aconsideration of the drawings and ensuing description of the invention.

SUMMARY OF THE INVENTION

The present invention is a method and apparatus for equalizing thefrequency response of a transmission line. The method includes the stepsof modeling the frequency response of the transmission line for apredetermined frequency range to a predetermined accuracy; determining adesired equalizer response by taking an inverse of the modelledfrequency response of the first step; implementing an equalizer thatexhibits the desired response; and utilizing the equalizer to equalizethe frequency response of the transmission line. The apparatus includesan adaptive equalizer circuit which includes a plurality of signalprocessor circuits which each take an input signal from the transmissionline and process it to mimic a term in a transfer function whichrepresents an inverse of the transfer function of the transmission line.The signals from these processors are then summed and multiplied by aprogrammable gain term. Then the input is added to the output of themultiplier to form an output equalizer signal. The programmable gainterm is adapted to cause the signal at the output to closely approximatea predetermined high frequency signal peak value through a send loopmechanism.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a representation of a transfer function H(s) where srepresents the complex frequency, z is the zero frequency (rad/sec) ofthe transfer function and p is the pole frequency (rad/sec) of thetransfer function above.

FIG. 1B is a plot showing the frequency response of the transferfunction H(s). The vertical axis is gain in db (20 loglH(jω)l), thehorizontal axis is angular frequency, ω, in radians per second. Thedashed curve is an asymptote of the transfer function H(s). The solidcurve is a plot of the transfer function H(s).

FIG. 2A is a representation of a transfer function E(s) where srepresents the complex frequency. H₁ (s) is a high pass transferfunction (shown as a second order function specifically here), ω_(n) isthe natural frequency of H₁ (s), ξ is the damping factor of H₁ (s), k isthe variable gain (between 0 and 1) fitted to the transmission line typeand length.

FIG. 2B is a plot of gain versus frequency for four curves A, B, C andD. Curve A is a constant unity gain function. Curve B is the frequencyresponse of H₁ (s). Curves C and D are two different equalizer responsecurves corresponding to gain value k₁ and k₂, respectively, where k₁>k₂.

FIG. 3 is a representation of a model of a communications systemcomprising a transmitter, a transmission line and a receiver.

FIG. 4A is a representation of a transfer function which is the inverseof a transmission line transfer function where a number of terms areutilized and summed to approximate the desired transfer function asshown. The R₀ term represents the characteristic impedance of thetransmission line, the R₁ through R_(N) terms represent attenuation dueto skin effect in the transmission line. The k term is proportional tothe length of the transmission line and ω₂ through ω_(n) are the set ofcorner frequencies representative of frequencies where the skin effectincreases in a particular model.

FIG. 4B is a plot of gain (dB) versus frequency (Hz) showing a pair ofcurves E and F. Curve E is a dashed line and represents the inverse ofthe measured frequency response of a 100 meter length of category 3 UTP,i.e., what the equalizer response should be. Curve F, the solid line,represents a modelled equalizer transfer function using a model as shownin FIG. 4A to closely approximate an ideal equalizer.

FIG. 5 is an electrical schematic diagram showing a low pass"tap"implementation according to a presently preferred embodiment of thepresent invention.

FIG. 6 is an electrical schematic diagram showing a servo loop forgenerating the control voltage applied to the gate of MOS transistor 64(FIG. 5) to bias the transistor to act like a precision resistor.

FIG. 7 is an electrical schematic diagram showing on the left an allpass function with a gain of "x" and on the right a low pass functionwith signal inversion corresponding to a transfer function of ##EQU1##where ω is 1/RC and s is the complex frequency. R is the value of theresistance of MOS transistor 64 and C is the value of the capacitancedenoted at 70.

FIG. 8 is a block diagram showing an equalizer apparatus implementingthe transfer function described in FIGS. 4A and 4B.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Those of ordinary skill in the art will realize that the followingdescription of the present invention is illustrative only and is notintended to be in any way limiting. Other embodiments of the inventionwill readily suggest themselves to such skilled persons from anexamination of the within disclosure.

The present invention is a method and apparatus for equalizing thefrequency response of a transmission line. The method includes the stepsof modeling the frequency response of the transmission media for apredetermined frequency range to a predetermined accuracy; determining adesired equalizer response by taking an inverse of the modelledfrequency response of the first step; implementing an equalizer thatexhibits the desired response; and utilizing the equalizer to equalizethe frequency response of the transmission line. The apparatus includesan adaptive equalizer circuit which includes a plurality of signalprocessor circuits which each take an input signal from the transmissionline and process it to mimic a term in a transfer function whichrepresents an inverse of the transfer function of the transmission line.The signals from these processors are then summed and multiplied by aprogrammable gain term. Then the input is added to the output of themultiplier to form an output equalizer signal. The programmable gainterm is adapted to cause the signal at the output to closely approximatea predetermined high frequency signal peak value through a send loopmechanism.

Transmission Line Model

Turning now to FIG. 3, a circuit diagram of a theoretical model of acommunications system 20 is shown. The communications system 20 is madeup of models of a transmitter 22, a transmission line 24, and a receiver26 connected in series. A signal to be communicated is generated by thetransmitter 22, carried by the transmission media 24, and received anddemodulated by the receiver 26.

The transmitter 22 is modeled as a voltage source 28 and a sourceresistance 30. The input voltage has a form of v_(i) (s). The sourceresistance 30 has a value of R₀.

The receiver 26 is modeled as a load resistor 32. The load resistance 32also has a value of R₀. As those of ordinary skill in the art willrealize, the source resistance 30 and the load resistance 32 aredesigned to be matched to reduce signal reflections. The output voltageof the system is the voltage drop across the load resistor 32 and hasthe form of v₀ (s).

As those of ordinary skill in the art will realize, cable loss in mostwired transmission media such as coax, UTP and STP results from a skineffect where electrical conduction is limited to the surface area,rather than the whole cross section, of the conductor at higherfrequencies. This causes the series impedance and thus the insertionloss to increase with frequency.

In order to properly model the transmission line frequency response, onemust take into account that the impedance of the transmission linechanges with frequency. In accordance with the present invention, thetransmission line frequency response is modelled with a plurality ofelements or terms. In FIG. 3, they are shown as a plurality of "taps"36, 42, 48, each including a resistance and inductance wired inparallel. In FIG. 8, five such terms are used to form an equalizer asshown. For a transmission line which will be used transmit signals overa given frequency range of interest, a number of "taps" will preferablybe used to approximate a series of frequency bands within the range ofinterest. The number of parallel taps and the values of the componentswithin each tap depend on the frequency range that is of interest andthe degree of accuracy that is desired. The number of parallel taps canvary from as few as one up to any number that is desired. In general, itis the case that, the greater the frequency range and the greater theaccuracy, the greater the number of taps required.

The transmission line 24 is modeled according to the present inventionas several elements in series. The first element is a resistance tap 34.This represents the zero frequency (i.e. DC) or ohmic loss of thetransmission line 24. The resistance tap 34 has a general value of kR₁where k is a scaling factor that is proportional to the length of thetransmission media 24.

The remaining elements in the model of the transmission line 24 areparallel taps. The first parallel tap 36 consists of resistor 38 andinductor 40 where the resistor 38 has a value of kR₂ and the inductor 40has a value of kL₂. The second parallel tap 42 consists of resistor 44and inductor 46 where the resistor 44 has a general value of kR₃ and theinductor 46 has a general value of kL₃. The final parallel tap 48 isdesignated as the nth parallel tap and consists of resistor 50 andinductor 52 where the resistor 50 has a general value of kR_(n) and theinductor 52 has a general value of kL_(n). As noted above, the number ofparallel taps can vary from as few as one up to any number that isdesirable. As a result, n can be any positive integer desirable.

The corner frequency of each tap 36, 42, and 48 is of the form ω_(c)=R/L while the insertion loss caused by each tap is of the form kR whichdepends on the length of the transmission line. Note that the cable lossfor a given frequency is modelled by the total insertion loss for all ofthe taps with corner frequencies lower than the given frequency.

The resulting transfer function which models the cable loss as afunction of frequency takes the form of ##EQU2## where ω₂ =R₂ /L₂, ω₃=R₃ /L₃, and ω_(n) =R_(n) /L_(n).

Transmission Line Equalizer Response

Based on this approximation, a cable equalizer has to exhibit theinverse of Equation 1 for proper compensation. The resulting transferfunction which models the cable equalizer as a function of frequencytakes the form of ##EQU3## where ω₂ =R₂ /L₂, ω₃ =R₃ /L₃, and ω_(n)=R_(n) /L_(n).

Those of ordinary skill in the art will recognize Equation 2 as an allpass function that is added to a high pass function. Also, each of thetaps 36, 42, and 48 are represented as high pass functions of the formRs/(s+ω_(c)). The overall transfer function can be implemented using asummation approach.

FIG. 4A is a representation of a transfer function which is the inverseof a transmission line transfer function where a number of terms areutilized and summed to approximate the desired transfer function asshown. The R₀ term represents the characteristic impedance of thetransmission line, the R₁ through R_(N) terms represent attenuation dueto skin effect in the transmission line. The k term is proportional tothe length of the transmission line and ω_(n) is the set of cornerfrequencies representative of frequencies where the skin effectincreases in a particular model.

FIG. 4B is a plot of gain (dB) versus frequency (Hz) showing a pair ofcurves E and F. Curve E is a dashed line and represents the inverse ofthe measured frequency response of a 100 meter length of category 3 UTP,i.e., what the equalizer response should be. Curve F, the solid line,represents a modelled equalizer transfer function using a model as shownin FIG. 4A to closely approximate an ideal equalizer.

Transmission Line Equalizer Design

Circuit diagrams showing a tap implementation according to the presentinvention are shown in FIGS. 5, 6 and 7. As those of ordinary skill inthe art will recognize, for tap frequencies in the megahertz and tens ofmegahertz range, it is desirable to process the signal to be compensatedin the current domain so that one can do away with bandwidth limitinghigh impedance nodes along the signal path. Consequently, this is thepreferred method of implementation.

Turning now to FIG. 5, a circuit diagram showing a low pass tap 60according to the present invention is shown. The low pass tap 60 has afirst MOS transistor 62 that has the drain tied to the gate and thesource tied to ground to form a diode. The gate of the first MOStransistor 62 is also tied to the source of a MOS resistor 64. The gateof the MOS resistor 64 is tied to a control voltage source 66 whichprovides a variable DC voltage to set the resistance of MOS resistor 64.The drain of the MOS resistor 64 is tied to the gate of a second MOStransistor 68. The source of the second MOS transistor 68 is tied toground.

The low pass tap 60 uses the inherent capacitance between the gate andthe source of the second MOS transistor 68 and the resistance of the MOSresistor (or transistor) 64 to determine the corner frequency which isof the general form ω_(c) =1/RC. If the second MOS transistor 68 doesnot have the correct capacitance value desired, a capacitor 70 (shown inphantom) can be added between the gate of the second MOS transistor 68and ground to adjust the capacitance.

The corner frequency of the low pass tap 60 can also be manipulated bychanging the resistance of the MOS resistor 64. The resistance of theMOS resistor 64 depends on the magnitude of the control voltage appliedto the gate and the aspect ratio of the resistor itself. So, changingthe resistance of the MOS resistor 64 can be done in either or both oftwo ways. First, the value of the control voltage source 66 can bevaried. An increase in the voltage results in a decrease in theresistance and vice versa. Second, the aspect ratio of the resistoritself could be changed. One simple method of accomplishing this in anintegrated circuit implementation is to fabricate a plurality ofidentical MOS resistors on a common substrate and connect them in seriesor parallel with on chip switching until the desired resistance isachieved.

A presently preferred method of changing the resistance of the MOSresistor 64 is outlined below with respect to FIG. 6.

The resulting transfer function of the low pass tap 60 takes thesimplified form of ##EQU4## where ω=1/RC.

Turning now to FIG. 6, a circuit diagram showing a servo loop for MOSresistor control according to the present invention is shown. The servoloop is used in place of the control voltage source 66 of FIG. 5 tocontrol the MOS resistor 64 of FIG. 5 and the output at node "A" is thedesired voltage on the gate of the MOS resistor 64.

Amplifier 80 forces node 82 and node 84 to have the same voltage. Afirst current source 86 has a value of V_(REF) /R_(INT). A secondcurrent source 88 has a value of V_(REF) /R_(EXT) where R_(EXT) is aprecision resistor (not shown). A first resistor 90 and a secondresistor 92 are connected in series and have a combined resistance valueof aR_(INT) where "a" is a constant. So, the MOS resistor 64 is biasedwith a voltage drop equal to (aR_(INT))(V_(REF) /R_(INT)) or aV_(REF).Since the current through MOS resistor 64 is V_(REF) /R_(EXT), theequivalent resistance of the MOS resistor 64 is (aV_(REF))/(V_(REF)/R_(EXT)) or aR_(EXT).

This servo loop allows the integrated circuit designer to choose thevalue of the MOS resistor 64 by changing the aspect ratio of the MOSresistor 64 and the value of the series resistors 90 and 92 and have theresulting value of the MOS resistor 64 (i.e. aR_(EXT)) depend directlyon R_(EXT) which can be tightly controlled. The manufacturing techniquesfor the MOS resistor 64 and the series resistors 90 and 92 are lessprecise and could result in unwanted variations in resistance values.

Turning now to FIG. 7, a circuit diagram showing a high pass tap 100according to a presently preferred embodiment of the present inventionis shown. The high pass tap 100 is formed by the summation of an allpass function 102 and the low pass function 60 (FIG. 5) with the inputcurrent polarity reversed.

The all pass function 102 consists of a first MOS transistor 104 thathas the drain tied to the gate and the source tied to ground as a diode.The gate of the first MOS transistor 104 is also tied to the gate of asecond MOS transistor 106. The source of the second MOS transistor 106is tied to ground as a current mirror.

The all pass function 102 and the low pass function 60 are combined bytying the drain of MOS transistor 106 to the drain of MOS transistor 68.The resulting transfer function of the high pass tap 100 takes the formof ##EQU5## where ω=1/RC.

Recall in the above discussion of the model of the transmission line 24of FIG. 3 and the resulting transmission line equalizer response givenin Equation 2 that each of the taps 36, 42, and 48 of FIG. 3 arerepresented as high pass functions in Equation 2 having the formRs/(s+ω_(c)). This is the same form as Equation 4. Therefore, the highpass tap 100 of FIG. 7 could be used as a circuit implementation of eachof the taps 36, 42, and 48 of FIG. 3 and Equation 2.

The transfer function of Equation 2 as shown in FIG. 4 can therefore beimplemented as a circuit by first forming the high pass function as acircuit with an output, multiplying the output by a programmable gain,and adding an all pass function to achieve an equalizer output. Thedesign can then be made adaptive by having the programmable gainselected based upon measured response of the transmission line.

According to the present invention, the high pass function of Equation 2as shown in FIG. 4A can be implemented as a circuit by connecting aresistance tap having a value equal to that of the ohmic loss of thetransmission media in series with a number of parallel taps containingthe high pass tap 100 of FIG. 7. The number of parallel taps and thevalues of the components within each tap depends on the frequency rangethat of interest and the degree of accuracy that is desired as discussedabove.

The resistance tap can be implemented in any number of ways that areknown to those of ordinary skill in the art. According to the presentinvention, the resistance tap is implemented as a circuit similar tothat of the all pass tap 102 of FIG. 7.

The all pass function and the summation circuit can be of ordinarydesign that is well known to those of ordinary skill in the art. For anadaptive design, the multiplication circuit can also be of ordinarydesign.

The selection of the value of the programmable gain can be accomplishedin any number of ways which are also well known to those of ordinaryskill in the art. The presently preferred method of selecting theprogrammable gain is outlined below with respect to FIG. 8 whichexhibits a transfer function of the form of the transmission lineequalizer transfer function of Equation 2 and can be used to compensatefor the frequency response of the transmission line.

Turning now to FIG. 8, a transfer function of a transmission lineequalizer according to the present invention is shown with a high passfunction that has four taps. In this case, the corner frequencies of thehigh pass function 110 have been chosen to be ω₂ =1/R₂ C₂ =2π(2×10⁶), ω₃=1/R₃ C₃ 2π(6×10⁶), ω₄ =1/R₄ C₄ =2π(18×10⁶), and ω₅ =1/R₅ C₅ =2π(54×10⁶)rad/sec.

These corner frequencies are one possible choice for a transmission lineequalizer for a Fast Ethernet LAN device referred to as 100BaseT4designed to IEEE Standard 802.3u-1995 Clause 23. Under this standard,the transmission media for each channel is a STP or UTP of Category 3 orbetter having a characteristic impedance of 100 Ω and a maximum lengthof 100 m. The frequency range of interest is from 0 to 25 MHz.

Also shown in block diagram form in FIG. 8 is one means of selecting thevalue of the programmable gain of the multiplier 112. Recall that thisis what makes the transmission line equalizer design adaptive. First,the signal at the equalizer output 114 is processed by a peak detector116. Then the peak detector 116 compares the signal peak against areference to determine if the signal peak is within a specified range ofthe reference. If the signal peak is within the specified range, thenthe transmission line equalizer is providing proper compensation and theprogrammable gain is left unchanged.

When the signal peak exceeds the specified range, the peak detector 116sends a signal to decrement an up/down counter 118. The up/down counter118 in turn decreases the programmable gain of the multiplier 112. Thisresults in an decreased amount of compensation from the transmissionline equalizer.

When the signal peak falls below the specified range, the peak detector116 sends a signal to increment an up/down counter 118. The up/downcounter 118 in turn increases the programmable gain of the multiplier112. This results in an increased amount of compensation from thetransmission line equalizer.

Through the repeated application of this process, the frequency responseof the transmission line equalizer can be adjusted to better match thatof the transmission media whatever the specific media used or the lengthof that media.

If the signal at the equalizer output contains data pulses that vary inwidth, then the means of selecting the value of the programmable gain ofthe multiplier 112 can be improved by detecting the peak level of onlythe narrow pulses rather than the absolute peak of all pulses.

While illustrative embodiments and applications of this invention havebeen shown and described, it would be apparent to those skilled in theart that many more modifications than have been mentioned above arepossible without departing from the inventive concepts set forth herein.The invention, therefore, is not to be limited except in the spirit ofthe appended claims.

What is claimed is:
 1. A cable equalizer device for compensating for the frequency response of a transmission media that has a finite length and that can be modeled as an ohmic loss in series with at least one parallel tap consisting of an inductor and a resistor in parallel and wherein the frequency response of the device is the inverse of the frequency response of the transmission media and the device has an input node and an output node, the device comprising:a high pass function means added to a first all pass function means wherein the high pass function means has an input and an output and comprises the following elements connected in series:an ohmic loss proportional to the length of the transmission media and at least one parallel tap consisting of a second all pass function means minus a low pass function means.
 2. The device of claim 1 further comprising a multiplication circuit which has variable magnitude that is proportional to the length of the transmission media connected in series with the output of the high pass function means.
 3. The device of claim 2 further comprising a multiplication circuit magnitude control loop circuit, said control loop circuit having a peak detector and an up/down counter connected in series, said control loop circuit connected between the output node of the device and the multiplication circuit.
 4. The device of claim 2 wherein the second all pass function means comprises:a current source having an input and an output wherein the input is tied to a positive voltage potential; a first MOS transistor having a drain, a gate, and a source; and a second MOS transistor having a drain, a gate, and a source wherein the drain of the first MOS transistor is tied to each of the gate of the first MOS transistor, the gate of the second MOS transistor, and the output of the current source and the source of the first MOS transistor and the source of the second MOS transistor are both tied to ground.
 5. The device of claim 4 wherein the low pass function means comprises:a current source having an input and an output wherein the input is tied to a positive voltage potential; a first MOS transistor having a drain, a gate, and a source wherein the source is tied to ground; a second MOS transistor having a drain, a gate, and a source wherein the source is tied to ground; a control voltage source having a positive terminal and a negative terminal wherein the negative terminal is tied to ground; and a MOS Resistor having a drain, a gate, and a source wherein the drain of the first MOS transistor is tied to each of the output of the current source, the gate of the first MOS transistor, and the source of the MOS Resistor, the gate of the MOS Resistor is tied to the positive terminal of the control voltage source, and the drain of the MOS Resistor is tied to the gate of the second MOS transistor.
 6. The device of claim 5 wherein the low pass function means further comprises a capacitor tied between the gate of the second MOS transistor and ground.
 7. The device of claim 5 wherein the MOS Resistor has a constant effective area and the control voltage source applies a select DC potential to achieve a select resistance value for the MOS Resistor.
 8. The device of claim 5 wherein the control voltage source applies a constant DC potential and the MOS Resistor has an effective area which is changeable to achieve a select resistance value for the MOS Resistor.
 9. The device of claim 8 wherein the effective area of the MOS Resistor is increased by placing a plurality of identical MOS Resistors in series.
 10. The device of claim 8 wherein the effective area of the MOS Resistor is decreased by placing a plurality of identical MOS Resistors in parallel. 